Magnetic resonance imaging (MRI) systems have revolutionized the field of medical diagnostics since 1982. A seldom considered key feature of these systems is that they also represent the first large-scale, commercial, profitable and successful application of low-temperature superconductivity (LTS) used in their high-field (0.3-5 Tesla) magnets. More then 12,000 whole-body MRI systems have been installed in hospiitals worldwide, providing a power base for Cryogenics which has not yet been exploited. Their price tag is still very high: $1-2 million. Therefore, most people do not yet have access to the benefits of this superb new diagnostic tool. An MRI scan still costs between $500 and $1200. Drastic cost reduction of such MRI systems is therefore desirable.
A new dimension has been added to this scenario by the invention and discovery of High-Temperature Superconductivity (HTS) by Bednorz and Miller (IBM Züirich) in 1986. This opens up new possibilities for the electronics of an MRI system in particular and for other systems in general. The key (potentially cryogenic) components of an MRI machine (excluding the digital computer) are (in the sequence of costs):                The superconducting whole-body magnet (0.3, 0.5, 0.75, 1.0, 1.5, 3, 4 Tesla).        Three gradient power amplifiers for the generation of the gradient magnetic fields enabling the spatial resolution of an MRI picture in the x, y, and z axes. The pulse power levels per axis are in the order of magnitude of 300 V, 100-200 A, 30-60 kVA.        The RF pulse power amplifier (1-20 kW, 13, 21, 42, 64, 126, 168 MHz).        RF receive coils using high-temperature superconductors plus cryo-preamplifiers.        Add-on: Cryo-cooled laser diodes for laser surgery [2] and hyperpolarized inert gas MR imaging (patent pending).        Equipment for cryo-surgery.        
There is room and opportunity for the implementation of size and cost reduction in all of these components by the application of a new concept: Cryogenic Power Conversion (CPC) [M1-M20]. This concept is based on the MOSFET device properties shown in FIGS. 1 and 2: The drastic reduction in the loss producing on-resistance by cryo-cooling down to 77 K, the temperature of liquid nitrogen. Thus, one can envision a complete Cryo-MRI system which would be much smaller than the existing ones because all the components could be integrated into the main magnet assembly. This vision is illustrated by FIGS. 3 and 4.
A start has been made with researching the feasibility of a CPC implementation of the three MRI gradient amplifiers, which require high-speed PWM switching. Commercial applications include cost- and size-reduced Magnetic Resonance Imaging systems (MRI) providing improved performance for low-cost health care, MRI mass screening, etc. especially in rural areas and for export. Miniaturized, small size, high-efficiency cryogenic power conversion will also find applications in airplanes, ships, buses, trains, etc., especially in combination with high-temperature superconducting components.
In summary, it is believed that Magnetic Resonance Imaging (MRI) machines using superconducting magnets are well suited for a first implementation and demonstration of this CPC concept since cryogenics is already available, the required power levels are high, the duty cycles are low and various other system components such as the RF amplifiers [M9], HTS-RF receive coils, circuits using laser diodes for laser surgery [2], hyperpolarized inert gas MR imaging, and cryo-surgery equipment) can also be miniaturized by cryo-cooling. It is also assumed that the switching losses in a high-frequency (200-1000 kHz) switch-mode circuit can basically be eliminated or drastically minimized by soft-switching techniques.
3. A look at the history of the past half century clearly shows that there is no better size, weight and, therefore, cost reduction technology available than that provided by the Semiconductor Revolution. For example, a computer with the calculating power available in today's lap-top computer required the space of an entire office room thirty years ago. Unfortunately, power electronics has not yet benefited to the full possible extent from the potential provided by this silicon integrated circuit technology as have the fields of computers and communications. The reason is that the power density in power semiconductor devices is still too high. The key problem in the field of power electronics is to remove the high dissipation power generated by the semiconductor switching devices. A better solution is to reduce and minimize that dissipation at the source. This can be done by cryogenically cooling certain suitable power semiconductor devices such as MOSFETs: Cryo-MOSFET!
Cryogenic Power Conversion (CPC) using Cryo-MOSFETs requires, of course, the removal of some heat, which is not free. There are two approaches to this problem. The first is to provide a cryo-cooler for each circuit or system to be cooled. This is analogous to having a separate electrical generator for every house, a scheme which was found to be uneconomical a century ago when large generator stations in the multi-gigawatt range where built. Insistence on the one cryo-cooler per CPC circuit approach will kill the CPC concept. The second approach is to search for, to develop and to implement systems where a centralized cooling agent such as LN2 is employed in order to provide the cooling for several CPC circuits. Cyro-coolers are still relatively expensive due to the non-existent mass market at the present time.
A system which requires the cooling of several high-power components is now the proposed Cryo-MRI machine, the object of this invention. It requires the cooling of, first, the superconducting magnet, second, third and fourth the 3 gradient amplifiers (30-60 kVA each) for the 3 axes x, y, z in the 3-dimensional space [6], and fifth, the RF pulse power amplifier (1-10 kW pulsed) [21, 23]. In addition, one can add more cryo-cooled components such as the gradient coils themselves [22], RF receive coils using thin-film HTS inductors [13, 24] and the equipment required for Cryo-Surgery and/or Laser-Surgery in the future. A conventional gas laser transmitter for laser surgery could cost as much as $ 20k-50k while a cryo-laser diode may be much cheaper. The light output of semiconductor light-emitting diodes (LED) and lasers also increases by an order of magnitude or more through cryo-cooling. GaAs laser diodes for surgery as replacement for expensive gas lasers are under development [2]. In summary, one can envision 5-9 components of an MRI system which could benefit from cryo-cooling. Then a centralized cooling system becomes economical. In the case of superconducting (HTS) gradient coils [22] one would have very low losses and the high reactive gradient energy can be recovered at each pulse cycle.
In addition, cryo-computers exhibiting high clock frequencies can be added to the set of cryo-cooled components.
An assumption for this project is that the MOSFET switching losses can be drastically reduced by suitable “soft-switching” techniques and the conduction losses by cryo-cooling. (R. DeDoncker: “The switching losses can be virtually eliminated” [9].)
In order to understand the CPC concept [7, 10, 12, 16] one has to study the properties of the Cryo-MOSFET and other required components.
4. The Cryo-MOSFET and the IGBT
High-efficiency power conversion circuits require fast, low-loss switches. From the series of available switches (GTOs, MCTs, SCRs, MTOs, IGBTs, Bipolar transistors, BITs) the MOSFET is the fastest. This is one of its great advantages.
It has been found [7] that commercially available, non-FREDFET [16], power MOSFETs in low-cost plastic packages (TO-247, TO-264 etc.) work well when immersed in liquid nitrogen at a temperature of 77 K, even if they were in no way designed for such a “cool” operation. FIG. 1 shows the measured on-resistance of a typical device as a function of the drain current at 300 K and 77 K: APT10026JN (APT-MOS technology, ratings: 1000 V, 33 A, 0.26 Ω). The on-resistance improvement factor is about F=15 at low drain currents. It would be higher (350-400 K) for a DC measurement and in actual operation with internal junction heating at 300 K. One can see the drastic reduction of the loss-producing on-resistance by cryo-cooling in addition to a remarkable increase in the current handling capability up to 100 A-125 A. The on-resistance can now be made as small as desirable by paralleling more MOSFETs, a feature not possible with minority carrier devices such as IGBTs, MCTs, GTOs, SCRs, etc. which exhibit an on-state threshold voltage of 1-2 V. (“Silicon is cheap!”).
The new COOL-MOS devices from Siemens and IRC have even lower on-resistance both at 300 K and 77 K [26]. In a 500 V device the improvement factor comparing conventional MOSFET at 300 K with a COOL-MOS at 77 K is about F=25.
These measurements demonstrate that the conduction losses of MOSFETs can be reduced by an order of magnitude through cryo-cooling. The reduction of dissipation power permits the elimination of large heat sinks and provides the possibility to place the MOSFET chips closer together. As a result power circuits can be miniaturized. The switching losses of MOSFETs are lower than those of other devices since they are fast switching and do not exhibit the so-called “tail current”. High-speed MR imaging requires switchmode gradient amplifiers with a relatively high power bandwidth (>20 kHz). Therefore, high PWM switching frequencies (>200 kHz) are needed which only MOSFETs can provide. IGBTs (Insulated Gate Bipolar Transistors) are too slow. This means that today the Cryo-MOSFET has no competitor for the application of MRI gradient amplifiers.
5. Design Philosophy for MRI Cryo-Gradient Amplifiers
The key technical objective of the design of Cryo-gradient amplifiers is the implementation of high-speed PWM switching (200-500 kHz). Gradient amplifiers should provide the following:                inductive coil current IL cannot change instantly, therefore the current flows now Higher gradient pulse currents for smaller field-of-view imaging (magnification);        Faster gradient rise times for increased imaging speed;        Dramatically reduced size, weight, and space requirements for the power system;        Higher energy efficiency, and therefore reduced air-conditioning load in the hospital,        Considerably reduced cost of equipment and operation.All these features can be implemented applying the concept of CPC.Circuit Topologies: The Half- and Full-Bridge        
The key element of most high power electronics is the so-called half- or full-bridge (2 half-bridges) circuit shown in FIG. 5. For successful cryogenic power conversion one has to minimize all the losses in order to compensate for the cooling cost. The CPC concept for MRI is based on the assumption that even at higher frequencies the switching losses can be more or less eliminated by suitable soft-switching techniques to be found and implemented.
The following losses have to be considered:                The conduction losses which are reduced by a factor 10-15 at low currents and more at high currents by cryo-cooling and paralleling.        The voltage/current overlap losses to be reduced by fast- and soft-switching.        The losses due to MOSFET drain-source output capacity discharge during turn-on.                    *The MOSFET source-drain diode reverse recovery losses eliminated or reduced by using the bi-directionality of the MOSFET (MOSFET commutation) or through the use of fast recovery diodes (FREDs).                        
The overlap and capacity discharge losses can be minimized in the half-bridge topology by a technique called zero-voltage-switching/clamped voltage (ZVS-CV), also known as quasi-resonant, pseudo-resonant or transition-resonant ZVS techniques. (Ref. [4] page 186-189). It is based on the assumption that the active devices can be switched fast compared to the voltage transition time. This condition can best be implemented with the fast-switching MOSFET. All other devices, IGBT, MCTs, GTO, etc. exhibit longer turn-off and turn-on times. In addition, the MOSFET has the required commutating diodes already built-in. Thus, at least at lower frequencies, no external diodes are required. The on-state source-drain diode losses can be reduced in the Cryo-MOSFET by turning it on (MOSFET commutation). This feature is due to the MOSFET bi-directionality (positive/negative current flow) not available in any other power device: A great advantage.
The ZVS-CV soft-switching technique can be explained as follows [4]. Let us assume the upper MOSFET Q1 (FIG. 5) is turned on carrying a filter coil current IL. Q2 is off. At time t1 Q1 is turned off fast compared to the total transition switching time, ideally in zero nanoseconds. Now, for a dead time Δt(=t2−t1) both switches are off. The through the capacitors C1 and C2 charging C1 and discharging C2. At the same time the center pole voltage Vp changes from ˜VB down to −0.7 V, turning on the lower diode D2. It now carries the current until after the dead-time Δt at time t2 the lower switch Q2 is turned on. The voltage transition is free of loss since no current flows through the MOSFETs or diodes: No overlap losses. With no load the filter capacity CR now acts as the voltage source and reverses the current direction. IL has a triangular shape at 50% duty cycle. Ideally, the dead time Δt should be made equal to the capacitor discharge time ΔtC which is given by:       Δ    ⁢                   ⁢          t      c        =                    (                              C            1                    +                      C            2                          )            ·              V        s                    I      L      
At the turn-on of switch Q2 the inductor current IL changes its direction and flows first through D2 and then through Q2. After half a cycle time the same soft-switching transition repeats itself at time t3 for the current flowing through Q2.
A problem occurs at large (or small) duty cycles if the current IL is large and does not change its polarity. In this case the (C1+C2) capacity charge and discharge mechanism does not work and soft-switching occurs only at one and not at both transitions of a full cycle. At the turn-off off of switch Q2 the positive current transfers now from the switch to the commutating diode and the reverse recovery problems of that diode enter the picture. Unfortunately, the fast recovery diode MOSFETs (FREDFETs) do not work at 77 K due to some gate channel related “carrier freeze-out” effect [16]. But the reverse recovery charge/time of a normal MOSFET decreases with cryo-cooling, even if not as much as one would have hoped [16]. Paralleling fast FRED-diodes does not help since the on-state voltage of the cryo-MOSFET source-drain diode is lower than that of the fast recovery diodes. Using the MOSFET source-drain diode for the commutation function works fine at low frequencies (<10 kHz) if one slows down the switching speed of the transistor with large gate resistors (10-100 μl). This is necessary because the fast recovery of the diode produces dangerous dV/dt slopes which could retrigger the MOSFET to be turned off. In other words, while the MOSFET itself can handle fast voltage transients (20 V/ns-100 V/ns), its source-drain diode, if used, limits that permissible dV/dt value to much lower numbers: 2-5 V/ns. Since its reverse recovery time is relatively large (500-700 ns), and therefore the related reverse recovery loss high, the diode limits efficient high frequency switching.
The OCIA or S-Topology (Stanley-Topology): A very elegant solution to the MOSFET/commutating diode dilemma has just recently (PESC-97) been proposed by Gerald Stanley and Ken Bradshaw from Techron, a division of Crown International, Inc. [8]. It separates the MOSFET switch and the diode commutation functions thus permitting the use of Cryo-MOSFETs as well as high-speed FRED-diodes together. This Stanley-topology or S-topology is shown in FIG. 6. “The interleaved four-quadrant PWM power stage results in a doubling of the output ripple frequency while greatly reducing the ripple current amplitude. Shoot-through currents are reduced to low di/dt events that are readily controlled allowing zero deadtime operation” [8]. Here the MOSFET does not work as commutator. A key feature is that the pulse width of control signal S1 of an OC (opposed current) half-bridge is changed in the opposite direction to that of S2. A cryo-germanium diode would be best for this topology.
Test Results:
Full-Bridge: A full-bridge circuit using 4×4 MOSFETs APT5010LVR (FB-1) has been tested. The dead time is 400 ns, the current capability at 77 K is 4×50 A=200 A. The 2-filter ferrite or air inductors were 18 μH each (stranded AWG10 wire). FIGS. 7 (a-c) show waveforms of pole voltage (half-bridge centerpoint) and quasi-resonant filter inductor current IL for 250, 400 and 500 kHz at a supply voltage of 350 V, 350 V and 250 V respectively measured with the power circuit immersed in LN2.
The triangular coil current was measured with a Pearson (415-494-6444) current monitor No. 2879 where 1 V=10 A. One can see that smooth switching waveforms were obtained up to 400-500 KHz with the drive circuitry developed. The main limitation is the increased power dissipation in the driver ICs (TC4422). The peak-to-peak coil currents were 25.2 A at 250 kHz, 15.6 A at. 400 kHz and 8.3 A at 500 kHz. At higher frequencies the coil current is decreased because of the reduced coil charging time. Then the current is no longer sufficient to provide a clean charge/discharge of the MOSFET output capacitors as required for ZVS-CV soft switching. The waveforms start to show distortions and ringing as seen in FIG. 7c. This is also illustrated by FIG. 7d where the waveforms for 400 kHz are compared for 350 and 120 V. Again, at the low voltage the current (5.2 APP) is not sufficient for a clean soft-switching operation as at 350 V (20 APP). Note the transition from low to high dv/dt and the RF oscillations at 120 V.
It is also interesting to see how the rise and fall times of the drain-source voltage transitions change with frequency from 92 ns at 250 kHz to 145 ns at 400 kHz and 224 ns at 500 kHz (FIGS. 7a-c). This increase is caused by the fact that the coil current decreases which means that a longer time is required for the discharge of the output capacitors. An MRI gradient coil was simulated with a 573 μH inductor having a DC resistance of 100 mΩ.
Test Results for the S-Topology at 77 K: The half-bridge S-topology (Stanley-topology of FIG. 6 with 4 paralleled APT5012LNR MOSFETs (500 V, 47 A, 0.12 Ω), 2 APT60D60B FRED-diodes (600 V, 60 A, <70 ns) and with 2 air coils of 18.6 μH each has been tested. The dead-time was 360 ns. As expected the MOSFET turn-off transition of the pole voltage is soft-switching and the FRED-diode turn-off transition is fast- and hard-switching. Clean 220 A current pulses at 310 V with a PWM switching frequency of 150 kHz have been generated at 77 K with this circuit into a 586 μH gradient coil or a 1.3Ωload resistor, respectively.
In summary, it has been shown that switchmode gradient amplifiers can be designed operating at high frequencies (100-600 kHz) and at cryogenic temperatures (77 K). A high-speed fiber-optic cryo-cooled MOSFET driver system was developed providing good (no-load) soft-switching performance of 4 MOSFET (4×50 A) switch assemblies operating at 77 K up to 600 kHz at a 350 V supply. The classical full-bridge circuit with 16 MOSFETs as well as the new OCIA- or S-Topology (16 MOSFETs and 8 FREDs) have been built and tested. Current pulses of more than 200 A were delivered into resistive (1.3Ω) and inductive (576 μH) loads at 77 K. Since cryo-cooling speeds up diodes and MOSFETs very good RF (or even microwave) layout techniques must be employed for reliable cryo-power circuits. (This work was funded by the National Science Foundation under NSF-SBIR Contract No. DMI-9796142).
6. The MRI Cryo-RF Amplifier
FIG. 8 shows the temperature dependence of the efficiency of a 6.8 MHz Class E RF amplifier at an output level of 1 kW. It demonstrates the usefulness of cryo-cooling for RF generation needed in MRI systems. Using such an RF amplifier one can design an RF transmitter as required for MRI employing drain voltage modulation. The Modulator amplifier can be a cryo-gradient amplifier. A block diagram is shown in FIG. 9.
7. The Cryo-Speed-Up Inverter.
High-speed MR imaging of moving body parts such as the heart, the lungs, etc, require a higher voltage (1000-2000 V) gradient amplifier system. FIG. 10 shows the block diagram of a so-called Speed-Up Inverter used for this function. This circuit can, of course, also be implemented in the form of a cryo-circuit.
8. Brief Summary of the Invention
A brief (partial) summary of the invention is shown in FIGS. 3, 4 and 11. It demonstrates how RF amplifiers, gradient amplifiers and the speed-up inverters SI 1-S13 are all placed into a single cryo-dewar operating at 77 K or another suitable temperature between 77K and 200 K.